Integrated dual charge pump power supply and RS-232 transmitter/receiver

ABSTRACT

A monolithic integrated circuit containing an inverting/non-inverting voltage doubler charge pump circuit is disclosed for converting a unipolar supply voltage to a bipolar supply voltage of a greater magnitude. The unipolar input voltage is placed across a first external transfer capacitor by a first set of MOS switches during a first time period. The unipolar input voltage source is placed in series with the first transfer capacitor and this series combination of voltages is placed across a first external reservoir capacitor by a second set of MOS switches during a second time period. The voltage appearing across the first external reservoir capacitor is placed on a second transfer capacitor during the first time period by a third set of MOS switches. The voltage across the second transfer capacitor is placed into a second external reservoir capacitor with its polarity inverted by a fourth set of MOS switches during the second time period. A dual-collector lateral junction transistor, formed during the conventional CMOS processing steps used to fabricate the MOS switches, is connected as a voltage clamp between a ground potential and the two bipolar DC output lines of the power supply circuit to assure correct start-up conditions for the circuit. Gain reduction devices are placed in the semiconductor substrate to collect minority carriers which would otherwise be injected into inherent parasitic four layer PNPN junction devices created as a result of the architecture of the circuit, to prevent latch-up of the four layer devices. In a preferred embodiment, an RS-232 receiver and transmitter are contained on the same monolithic integrated circuit as the dual charge pump power supply.

This is a continuation of application Ser. No. 07/013,648, filed May 12,1987, now U.S. Pat. No. 4,809.152.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention pertains to charge pump power supplies forgenerating bipolar output voltages greater in magnitude than a singleunipolar input voltage. More particularly, the present inventionpertains to the integration of such a circuit on a single piece ofsemiconductor substrate material. Further, the invention pertains toother circuitry integrable on a single piece of semiconductor substratematerial along with such a power supply circuit.

2. Prior Art

Discrete component voltage doubler and voltage inverter circuits arewell known in the art. Such circuits are used in many electronic systemswhich require a multiplicity of DC voltages for operation. Morerecently, in the context of digital circuits and systems, it has becomecommon to employ a single five volt unipolar voltage supply to powerdigital circuitry in modern data processing systems. For example,semiconductor microprocessors, memories, and logic all commonly operatefrom a single five volt power supply. There are however, certaininterface circuits and other special purpose circuits which requirevoltages other than five volts. More particularly, some circuits requirevoltages in the ranges of from five to fifteen volts. Additionally,requirements often exist for bipolar power supply voltages so thatvoltage power requirements of plus or minus 15 volts and plus or minus12 volts are commonly encountered, for example in RS-232 communicationloops.

For these communication circuits and other applications, bipolar DCpower requirements are low when compared to the digital circuitry powerrequirements. In fact, it is common to encounter five volt unipolarpower supplies for driving digital logic rated in tens or hundreds ofwatts whereas interface and other power requirements may be as low astens or hundreds of milliwatts.

It is therefore often desirable to generate locally the variousnon-primary voltage sources, i.e., the bipolar voltage sources, if thepower requirements are not high and if it can be done economically andwith relatively high electrical power conversion efficiencies.

For an example, a minicomputer may have a 100 watt 5 volt power supplywhich supplies all of the requirements for a multiplicity of printedcircuit boards holding logic integrated circuits. On one of thoseintegrated circuit boards, there will often be an RS-232 digitalinterface circuit requiring a plus or minus 10 or plus or minus 15 voltpower supply. This interface circuit may consume 50 milliwatts of power.Instead of generating the plus and minus 15 volt power supply from themain power supply and then bussing these voltages to the boards whichrequire them, it is often more economical to generate these two voltagesfrom the bussed five volt power supply locally on whatever board needsother voltages. However, generating such voltages by the use of discretecomponents is often disadvantageous because the additional componentsrequired to generate such voltages take up a relatively large amount ofcircuit board real estate, and often are power inefficient, i.e., heatproducing.

The industry has recently turned its attention to attempts to furnishauxillary power supplies of the nature herein described on a singlesemiconductor substrate. Such circuits have the obvious advantages ofspace saving, assembly labor savings, and relatively lower powerdissipation. A form of such circuits known as charge pumps have beenused in semiconductor memory chips to produce a crude back bias supplyand for supplying the higher voltages needed to program such memorydevices. Charge pump circuits have been used in the inverting mode toproduce voltage polarities opposite to that of the supply voltage fromwhich they are generated. An example of such a circuit is found in theproduct designated ICL 7660, a power supply circuit manufactured by theassignee of the present invention.

The efforts to design and implement a bipolar charge pump integratedcircuit have met with several obstacles which result from the inherentnature of the integration process and the fabrication process which areused to manufacture these devices. It is well known to those in the artthat when MOS or CMOS circuits are integrated onto a singlesemiconductor substrate, the chip layout geometry and architectureinherently produce parasitic junction devices. These devices includejunction biodes, bipolar transistors, and PNPN four layer diode devices,similar to silicon controlled rectifier (SCR) devices.

The existence of these parasitic devices has created difficulties in thedesign and fabrication of dual polarity charge pump power supplycircuits. When forward biased, the aforementioned four layer diodedevice will cause a CMOS circuit to experience a phenomenon known aslatch-up. Latch-up is a phenomenon common to CMOS circuits whereby thecircuit can be triggered into a low impedance conducting state byforward biasing an inherent four layer diode device in the circuit. Thisfour layer diode may be triggered by various means into a low voltage,low impedance state. When this occurs, operation of the circuit isinhibited and possible damage may occur to the circuit if there is noinherent current limiting designed into the circuit.

Another problem inherent in the design of dual polarity charge pumpinverter circuits is the difficulty of assuring correct start-up of thecircuit. The conditions existing in the semiconductor material at thetime of start-up may randomly produce states which prevent such acircuit from ever starting up to produce the desired output voltages. Inthe past, elaborate systems and considerable extra circuitry has beendesigned into such circuits in an attempt to avoid this problem.

BRIEF DESCRIPTION OF THE INVENTION

The present invention consists of a CMOS inverting and non-invertingcharge pump power supply integrated into a single piece of semiconductorsubstrate material. An inherent lateral bipolar transistor formed duringthe CMOS fabrication process is utilized to always assure the correctoperating conditions which will allow start-up of the circuit. Inaddition, the inherent four layer diode devices which are created duringthe fabrication of the circuit are identified during the geometry layoutprocess which defines the locations on the semiconductor substrate wherethe various devices will be placed, and extra minority charge collectorregions are placed in the semiconductor substrate to collect injectedminority charge carriers and prevent the possibility of triggering theinherent four layer PNPN junction into a low impedance conducting orlatch-up mode.

Another aspect of the present invention is the integration, on a singlepiece of semiconductor substrate material, of an inverting charge pumppower supply, a non-inverting charge pump power supply, and acombination of RS-232C receivers and transmitters. Combination ofRS-232C transmitters and receivers may consist of at least onetransmitter together with either zero or any number of receivers.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1(a) is a simplified schematic diagram of the charge pump circuitof a preferred embodiment of the present invention.

FIG. 1(b) is a schematic diagram of the charge pump circuit of FIG. 1(a)wherein the switches are replaced by MOS transistors.

FIG. 2 is a gate drive circuit suitable for operating driving the gatesof the charge pump circuit of FIG. 1(b).

FIG. 3 is a schematic diagram of a preferred embodiment of the presentinvention further showing the substrate connection of the MOS devicesand a PNP lateral junction device for assuring the correct start-upconditions of the charge pump circuit.

FIG. 4 is a semiconductor substrate profile drawing of NPN lateraltransistor suitable for use in the present invention.

FIGS. 5(a) and 5(b) are respectively a schematic representation of afour layer device and a semiconductor substrate profile drawing of sucha device showing the MOS geometry which inherently creates such adevice.

FIG. 6(a) is a schematic diagram of a four layer device having extra Pregion collectors, suitable for use in the present invention.

FIG. 6(b) is a substrate profile drawing of a four layer device suitablefor use in the present invention having extra minority charge carriercollectors for preventing latch-up showing the relative placement ofsuch charge collectors.

FIG. 7 is a block diagram of an embodiment of the present inventionincluding a dual integrated charge pump power supply and a RS-232Creceiver and transmitter.

DESCRIPTION OF A PREFERRED EMBODIMENT

Referring first to FIG. 1(a), simplified conceptual schematic drawing ofthe basic charge pump circuit of the present invention, the circuit ofthe present invention operates by placing an input voltage upon one ofthe two transfer capacitors via a series of switches. The charge in thatcapacitor is then transferred to one of two reservoir capacitors. Thepolarity of the voltage is established via the switch interconnectingscheme.

More specifically, the operation of the circuit of FIG. 1(a) istime-divided into two segments, or phases. In a first phase, voltagefrom a voltage source is placed on transfer capacitors. During a secondphase the voltage on the transfer capacitors is transferred to thereservoirs capacitors.

Referring first to the positive voltage doubler portion of the circuit,transfer capacitor 10 is charged from voltage source 12 (having a valueVcc) by closing switches 14 and 16 while switches 18 and 20 remain openduring a first phase. During a second phase switches 14 and 16 are openand switches 18 and 20 are closed.

As can be seen from FIG. 1(a), when switches 18 and 20 are closed duringthe second phase the voltage source 12 is effectively placed in serieswith the voltage stored across reservoir capacitor 10 and thus the sumof the voltage across voltage source 12 and capacitor 10 is placedacross reservoir capacitor 22.

The inverting portion of the voltage doubler circuit operates asfollows: transfer capacitor 24 is charged to the voltage acrossreservoir capacitor 22 via the switches 26 and 23 which are closedduring the first phase of operation of the circuit while switches 30 and32 remain open. During the second phase of circuit operation switches 26and 23 are opened and the voltage across transfer capacitor 24 is placedon reservoir capacitor 34 via the closing of switches 30 and 32. Thoseof ordinary skill in the art will note that the circuit configuration issuch that when the voltage across transfer capacitor 24 is placed acrossreservoir capacitor 34 the positive end of transfer capacitor 24 isconnected to ground line 36 through switch 32 and the negative end ofcapacitor 24 is connected to the side of reservoir capacitor 34connected to -2 Vcc output line 38. The polarity of the voltage acrossreservoir capacitor 34 with respect to ground line 36 is thus such thatthe voltage across reservoir capacitor 34 is negative. The output ofreservoir capacitor 22 is connected to +2 Vcc output line 40.

The first and second phases of circuit operation described above arerepeated at a frequency which may range from approximately 100 hertz to100's of kilohertz or higher. It has been found that a frequency ofapproximately 15 K Hz performs satisfactorily for the purposes of thepresent invention.

The foregoing represents an idealized characterization of the operationof the circuit of FIG. 1(a). Those of ordinary skill in the art willreadily realize that it will take several first and second phase cyclesbefore the resultant voltage between ground terminal 36 and +2 Vccoutput terminal 40 actually reaches a voltage value of +2 Vcc. Likewise,it will be appreciated that several cycles are also needed for thevoltage between ground terminal 36 and -2 Vcc output terminal 38 arrivesat a voltage of -2 Vcc.

Those of ordinary skill in the art will also realize that the amount ofcurrent which may be drawn from the output of the circuit of FIG. 1(a)depends on the relative sizes of transfer capacitors 10 and 24 andreservoir capacitors 22 and 34, as well as the on impedance of switches14, 16, 18, 20, 26, 28, 32 and 30.

It will also be apparent that the voltage appearing between outputterminals 36 and 38 or 36 and 40 will be approximately twice the inputvoltage supplied by voltage source 12. Those of ordinary skill in theart will appreciate that other multiples of the input voltage Vcc atvoltage source 12, are readily achievable using the concept of thepresent invention.

Turning now to FIG. 1(b), it is seen that in an actual embodiment of thepresent invention switches 14, 16, 18, 20, 26, 28, 30 and 32 have beenreplaced with MOS transistors. Thus, switch 14 is replaced by P-channelMOS transistor 14 (a), switch 16 is replaced by N-channel MOS transistor16(a), switch 18 is replaced by P-channel MOS transistor 18(a), switch20 is replaced by P-channel MOS transistor 20(a), switch 26 is replacedby P-channel MOS transistor 26(a), switch 28 is replaced by N-channelMOS transistor 28(a), switch 30 is replaced by N-channel MOS transistor30(a), switch 32 is replaced by N-channel MOS transistor 32(a).

The time controlled operation of the circuit of FIG. 1(b) is implementedby phase control unit 42. Phase control unit 42 drives all of the gatesof the MOS devices, 14(a), 16(a), 18(a), 20(a), 26(a), 28(a), 30(a), and32(a) via gate control lines 44 and 46. Gate control lines 44 and 46 areconnected to the gates of the P-channel MOS transistors and N-channelMOS transistors, used as MOS switches, such that the switches are turnedon and off appropriately as described herein during the first and secondphases of the circuit operation.

Those of ordinary skill in the art will readily appreciate that, inorder to insure efficient power transfer, that switching of MOSswitching devices should be accomplished on a break before make basisor, at worst case, on a simultaneous switching basis. Those of ordinaryskill in the art will also realize that the order of phases could bereversed. Alternatively, a first clock could be used to control the setsof MOS switches controlling transfer capacitor 10 and reservoircapacitor 22, and a second clock could be used to control the sets ofMOS switches controlling transfer capacitor 24 and reservoir capacitor34.

It should be understood for purposes of this disclosure, that all of thecapacitors shown in FIGS. 1(a) and 1(b) would be located outboard of theintegrated circuit in an actual embodiment. That is, these capacitorsare external components which connect to the integrated MOS switches onthe semiconductor substrate via terminals provided on the semiconductorsubstrate for that purpose. For an operating frequency of 15 KHz, 20microfarads is a sufficient size for all capacitors. Those of ordinaryskill in the art will readily appreciate that as the switching frequencyis increased the values of the capacitors will drop, but that switchinglosses will increase due to the changing to discharging at the clockrate of its parasitic nodal capacitances in the MOS devices. Conversely,as the switching frequency decreases, the size of the capacitors wouldincrease, with the concomitant disadvantage that the increasingcapacitor size is accompanied by increasing physical size of thecapacitors.

For a current capacity of 10 milliampere at +10 volts and -10 volts, theMOS switching devices should have a channel width to channel lengthratio of 5000 to 10,000, with channel lengths of approximately fivemicrons. Those of ordinary skill in the art will recognize that therange of current output of the circuit described herein could be aslarge as approximately one ampere, however, the MOS devices would haveto be scaled accordingly as is well known in the art.

Referring now to FIG. 2, an embodiment of the phase control unit 42 ofthe present invention, the operation of phase control unit 42 isdisclosed. Those of ordinary skill in the art will recognize that phasecontrol unit 42 may consist of three conventional CMOS inverter circuitseach comprised of a P-channel and N-channel MOS transistor pair. Theembodiment of FIG. 2 contains a first CMOS inverter comprised ofP-channel MOS transistor 44 and N-channel MOS transistor 46, theinverter comprised of P-channel MOS transistor 48 and N-channel MOStransistor 50, and the inverter comprised of P-channel MOS transistor 52and N-channel MOS transistor 54.

These three inverter pairs are driven by oscillator 56, which may be anyconventional oscillator configured from CMOS elements as is well knownto those skilled in the art.

The circuit of FIG. 2 is powered by +2 Vcc and -2 Vcc lines 36 and 40.This assured that the voltage swing on gate lines 44 and 46 will spanapproximately the entire power supply range, thus ensuring that all ofthe gates of the P-channel and N-channel devices which they drive willbe as fully turned on as possible and can be turned off since alltransistors and enhancement types. This will guarantee as low anon-state impedance of the MOS switches as possible thus maximizing theefficiency and current drive capabilities of the present invention.

In the illustration of a preferred embodiment of the present inventiondepicted in FIG. 1(a), the substrate connections of the MOS devices areshown uncommitted. Those of ordinary skill in the art will realize thatjunction isolated MOS transistors such as used in FIG. 1(b) arefour-terminal devices and that both the gate terminal and the substrateterminal are control terminals. The turn-on voltage of the gate terminalis affected by the reverse bias on the substrate to source junction. Asthat reverse bias is increased, the turn on voltage of the device alsoincreases. The affect is significantly greater for an N-channeltransistor than a P-channel transistor.

As the substrate-to-source voltage increases the gate turn voltage ofthe device also increases, thus potentially increasing the on resistanceof the device to a point where circuit operation could be seriouslyeffected. Since, in a circuit of this nature, the drain-sourceresistance in the on state should be as low as possible, it is desirableto connect each N-channel MOS transistor substrate to its source.

With respect to P-channel transistors the effect of this reversesubstrate source biasing is about half of that for N-channel MOStransistors due to lighter channel impurity doping densities. The mostpractical solution in the case of the P-channel MOS transistors is toconnect all P-channel substrates to the most positive voltage in thecircuit. That voltage is, as seen from FIG. 1(a) +2 Vcc which appears onpositive supply line 40. These connections are shown in respect to FIG.3.

Prior to start-up, it is reasonable to assume that zero voltage existson all capacitors. At start-up, reservoir capacitor 22 may be connectedto ground line 36 or to -2 Vcc line 38. Reservoir capacitor 22 will beimmediately charged with the source substrate diodes of P-channel MOStransistors 14(a) and 18(a) to a voltage of approximately Vcc -0.6 of avolt. The voltage on reservoir capacitor 34 could lie somewhere betweenground line 36 and the voltage on reservoir capacitor 22; dependingwhich of transistors 26(a), 28(a), 30(a) or 32(a) were conducting (ifany). This results in a voltage on the -2 Vcc line that could be suchthat N-channel that transistor 16(a) and other transistors being turnedon. Under these conditions, a voltage between +2 Vcc drives the gates ofall of the output transistors and is indeterminate. Thus both start-upand operation is not assured.

If the other possible start-up conditions of the capacitor and MOSdevice connections and off/on states are assumed, those of ordinaryskill in the art will readily appreciate that the start-up and operationof the circuit of FIGS. 1 and 2 is not assured.

The solution to this dilemma is to place a clamp on the -2 Vcc line 38to clamp that voltage line to assure that it will never assume a voltagesubstantially more positive than that appearing on ground line 36. The+2 Vcc line 40 is also clamped so that it will never assume a voltagesubstantially more negative than the voltage Vcc on approximately Vcc-0.6 volts.

While those of ordinary skill in the art will realize that,conceptually, a diode would be an ideal clamping means for the -2 Vccline 38, it is not possible to fabricate a simple PN junction diode in aMOS process. A junction transistor will always be created by an attemptto fabricate a diode. The presence of such a transistor in the circuitof FIG. 1(b) would cause excess wasted current to flow in the circuit,because of its beta or current gain.

In a preferred embodiment of the present invention, this clamp iscomprised of a lateral NPN transistor. This lateral NPN transistor isshown in FIG. 3. The lateral collector and base of this device are bothconnected to -2 Vcc line 38, its vertical collector connected to +2 Vcc.The lateral collector serves to minimize the effective current gain ofthe unwanted but inherent vertical collector of NPN transistor 58, whichwould otherwise cause excess current flow from the +2 Vcc line throughground. Unless the -2 Vcc line 38 exceeds ground by approximately 0.6 ofa volt in the positive direction, this device will not conduct current.If the -2 Vcc line equals approximately 0.6 volts, the device turns onand current will flow in approximately equal portions through bothcollectors to maintain the -2 Vcc line at no greater than zero plusapproximately 0.6 volts.

With respect to the clamp for +2 Vcc line 40, the action of the inherentjunction diodes 59(a) and 59(b) present between the drain and substrateof devices 14(a) and 18(a) serve to clamp the +2 Vcc line to a voltageno more negative than the input positive supply voltage Vcc minusapproximately 0.6 volts.

Consequently the voltages on +2 Vcc line and -2 Vcc line are both welldefined. Additionally the voltage difference between +2 Vcc line 38 and-2 Vcc line 40 at start-up is (Vcc -1.2) volts and is also well defined.This value of voltage is sufficiently large to guarantee operation ofthe drive circuitry for the gates of the output transistors until thecharge pumps charge the lines +2 Vcc (40) and -2 Vcc (38) to thosevoltages.

The lateral NPN transistor used to clamp -2 Vcc line 38 is fabricatedusing conventional CMOS fabrication techniques. For a current drain ofplus and minus 10 mA at 10 volts, the periphery of the emitter for thelateral NPN transistor can typically be 100 microns. Those of ordinaryskill in the art will appreciate that the size of this device may bescaled to accommodate larger current carrying requirements, and itsperiphery need not be larger than 1000 microns.

Referring now to FIG. 4, a substrate profile drawing of a dual collectorlateral NPN transistor 58, that transistor 58 is fabricated on a portionof the lightly doped N type substrate material 60 in a P-well 62. P-well62 is placed into substrate 60 using common CMOS processing techniques.N region 64 serves as the emitter of the lateral NPN transistor, and issurrounded by N region 66 which serves as the lateral collector. Pregion 68 in P-well 62 serves as the base contact, it being understoodby those skilled in the art that P-well 62 itself serves as the base oflateral NPN transistor 58. N-region 70 located in a region of substrate60 outside of P-well 62 serves as the unwanted, but inherent verticalcollector of NPN lateral transistor 58.

When the base emitter junction of lateral NPN transistor 58 is forwardbiased, minority carriers injected by the emitter into the base arecollected by both the vertical and lateral collectors in roughly equalamounts. Connecting the lateral collectors to the common base reducesthe vertical collector current to approximately 1/2 of the clampcurrent. If a vertical NPN transistor had been used alone, the clampcurrent (base current) would be multiplied by the beta (approximately500 of the device) thereby wasting large amounts of current.

Referring now to FIGS. 1(b) and 3, during start-up reservoir capacitor22 is charged by the forward biased condition of the source-substratediodes 59(a) and 59(b) of P-channel device 14(a) and the drain substratediode of P-channel device 18(a). The initial current surge through thesediodes can be hundreds of milliamperes and thus be well above theholding current of the inherent SCR type four layer diode device whichexists in the circuit.

Such a four layer device is schematically represented in FIG. 5(a).Referring to FIG. 5(a), it is seen that the four layer device is made upof PNP transistor 72, NPN transistor 74, resistor 76, and resistor 78.Resistor 76 is connected across the base-emitter junction of PNPtransistor 72 while resistor 78 is connected across the base-emitterjunction of PNP transistor 74. The base of NPN transistor 74 isconnected to the collector of PNP transistor 72 and the base of PNPtransistor 72 is connected to the collector of NPN transistor 74. Theconnection of the emitter junction of PNP transistor 72 and resistor 76form the anode connection 78 of the four layer device and theintersection of resistor 78 and the emitter of NPN transistor 74 formthe cathode connection 80 of the four layer device.

As will be appreciated by those of ordinary skill in the art, the fourlayer device shown in FIG. 5(a) will enter a low impedance state betweenits anode 80 and cathode 82 after suitable triggering if the product ofthe betas of the two equivalent transistors is greater than one and theanode current into the four layer device is greater than the turn onvoltage of either transistor divided by its equivalent base emittershunting resistor, whichever is greatest.

Referring now to FIGS. 3, 5(a) and 5(b), it will be apparent to those ofordinary skill in the art that such a four layer device occurs in thecircuit of FIG. 3. The sources of either of P-channel devices 14(a) and18(a) (shown diagrammatically as P region 84 in FIG. 5(b)) represent theemitter of PNP transistor 72 of FIG. 5. The semiconductor substrate 60forms the base of PNP transistor 72 as well as the collector of NPNtransistor 74. P-well 86 forms the collector of PNP transistor 72 aswell as the base of NPN transistor 74. The source of either of N-channeltransistors 16(a) and 32(a), one of which is shown as N region 88 ofFIG. 5(b) forms the emitter of NPN transistor 74. Resistor 76 is formedby the bulk resistance of the P-well 86. Likewise, the resistor 78 isformed by the bulk resistance of the substrate material. Those skilledin the art will note that regions such as P region 90 in P-well 86 and Nregion 92 in substrate 60 serve as low resistance surface planescommonly used in CMOS technology to buss supply voltages to the surfacesof substrate and P-wells.

In order to trigger the four layer device into its low impedance state,currents must be injected into the base of either of transistors 72 or74, either the P-well 86 or the substrate 60. These currents must begreater than the holding current required for the four layer device.This condition can occur by various means. For example, a very rapidrate of increase in the anode-cathode voltage will force current intothe bases of transistors 72 and 74 due to the charging of thecollector-base junction capacitors inherent in those devices.Alternatively, forward biasing of a region in the substrate junctionadjacent to the P-well 86 and P-region 84 forming the emitter oftransistor 72 could induce base currents to flow in transistor 72 and 74sufficient to exceed holding current values. Either of these conditionscould occur at start-up of the circuit of FIG. 3.

In order for the inverting doubler charge pump circuit of the presentinvention to reliably operate, it is necessary to assure that thispossible latch-up condition can never occur. One method which is used insome CMOS circuits to inhibit the possibility of latch-up would be toinsert high value resistors in series with either or both of theemitters of NPN transistor 74 or PNP transistor 72. This method,however, in the present invention would result in an unacceptably highvalue of on impedance for the MOS switches.

Another method of assuring that the latch-up condition never occurs isdisclosed as an aspect of the present invention. The product of thebetas of PNP transistor 72 and NPN transistor 74 is mode less thanunity. Thus, the current flowing between the anode terminal 80 andcathode terminal 82 of the four layer device will never reach a valuegreat enough to equal the holding current necessary to sustain thatdevice in its low impedance state.

Referring now to FIG. 6(a), another four layer device 100 composed ofequivalent NPN and PNP transistors is shown. However, unlike the circuitof FIG. 5(a) the four layer device depicted in FIG. 6(a), having anodeterminal 102 and cathode terminal 104, a single collector NPN transistor106 and a multiple collector PNP transistor 108 as well as resistors 110and 112. The multiple PNP collectors (shown at 114) are tied back to thebase of the PNP transistor 108. Only a single multiple collector isconnected to the base of NPN transistor 106. These collectors 114 arefabricated on substrate 60 in a region located between the emitter ofNPN transistor 106 and the base of PNP transistor 108.

The function of the serial collectors 114 is to guard the forward biasedPN junction formed between P regions 128 or 138 and substrate 60 bycollecting the minority carriers which are injected into the substrate60. These carriers are thus prevented from reaching the base of the PNPtransistor 108 and assure that the beta product of these two transistorsis less than unity. Most of the minority carriers injected into thesubstrate are collected by these serial collectors before they candiffuse and be collected by the P-well which is also the base of the NPNtransistor. This may be designed to reduce the PNP beta to a value ofless than the reciprocal of the NPN beta thereby preventing latch-up.

Referring now to FIG. 6(b), a semiconductor profile drawing of fourlayer device 100 of FIG. 6(a), NPN transistor 106 is formed in P-well120. Contact 122, contacting N region 124 in P-well 120, constitutescathode 104 of four layer device 100. This N region may be the source ofeither N-channel MOS transistor 16(a) or N-channel MOS transistor 32(a)from FIGS. 1 and 3. N region 124 forms the emitter of NPN transistor 106and P-well 120 forms the base of NPN transistor 106. Substrate 60 formsthe collector of NPN transistor 106, as well as the base of PNPtransistor 112.

Contact 126, contacting P region 128 is at Vcc potential. P region 128may be either the source of P channel MOS transistor 14(a) or the drainof P-channel MOS transistor 18(a) from FIGS. 1 and 3. P region 128 formsthe emitter of PNP transistor 108.

P regions 130(a) through 130(e), in substrate 60, form the multiplecollectors of PNP transistors 108 (shown at 114 in FIG. 6(a)). Multiplecollectors 130(a) through 130(e) are connected together at the surfaceof the semiconductor substrate 60 by layer 132 which may be made ofaluminum and fabricated during the metalization step of a conventionalCMOS fabrication process. N regions 134(a) through 134(d), disposed inbetween P regions 130(a) through 130(e) are used for the purpose ofmaking a low impedance contact between the +2 Vcc line and thesubstrate. P-well 120, the base of NPN transistor 106, also serves asthe single collector of PNP transistor 108, as shown in FIG. 6(a). Theregions 135, shown adjacent to layer 132, are the gate oxide layer ofthe MOS structures.

As is shown in FIG. 6(b) the multiple collectors of PNP transistor 114are interposed in between the N-channel MOS transistor 16(a) in theP-well formed of N regions 124 and 136. This device, for illustration,shown as 16(a) on FIG. 6(b), has drain region 138 and gate 140. Thisdevice, for illustration, shown as 18(a) on FIG. 6(b), has drain region138 and gate region 142. P-channel MOS transistor 14(a) formed of Pregion 128 and P region 138. In this manner these multiple collectors130(a) through 130(e) are in a position to collect most of the minoritycarriers which are injected into the semiconductor substrate as a resultof forward biasing at start-up of the parasitic PN junctions formedduring the CMOS fabrication process.

Depending on the CMOS process used, the number of multiple collectors114 may range from 1 to approximately 10. Furthermore, the spacingbetween the injecting PN junction and the nearest P-well should betypically anywhere from 25 to 500 microns. Spacing may be reduced if thelifetime of the substrate minority carriers is particularly low and orthe substrate resistivity is very low (less than one ohm-centimeter). Inthe presently preferred embodiment the spacing between the injecting PNjunctions and the nearest P-well is approximately 150 microns and fourmultiple collectors 114 are used. This is based upon a process using asubstrate having a substrate resistivity of approximately 2.5 ohm-cm.

Although the presently preferred embodiment has been disclosed as aP-well CMOS embodiment, those of ordinary skill in the art willrecognize that N-well CMOS technology could also be used withoutdeparting from the spirit and scope of the present invention. Those ofordinary skill in the art will readily understand from the disclosureherein how to fabricate such an N-well embodiment.

Referring now to FIG. 7, a block diagram of a preferred embodiment ofthe present invention including dual charge pump power supply 200,previously described, RS-232C transmitter circuit 202, and RS-232receiver circuit 204. These elements are show diagrammatically asfabricated on a single piece of semiconductor substrate material 206.Positive reservoir capacitor 22 is shown connected to the semiconductorsubstrate via terminal pads 208 and 210. Negative reservoir capacitor 34is shown connected to the substrate via terminal pads 208 and 212.Positive transfer capacitor 10 and negative transfer capacitor 24 areshown connected to the substrate via terminal pads 214, 216, 218 and 220respectively. An input voltage is provided to the circuit at Vcc inputterminal pad 222 and ground input terminal pad 224. Those of ordinaryskill in the art will readily realize that ground input terminal 224 andterminal pad 208 may in some embodiments be the same connection terminalpad. The data input to RS-232 transmitter 202 is provided at terminalpad 226 and the output of RS-232 transmitter 202 is provided at terminalpad 228. The data input to RS-232 receiver 204 is provided at terminalpad 230 and the data output of RS-232 receiver 204 so provided atterminal pad 232.

A monolithic integrated circuit containing the dual charge pump powersupply 200 and RS-232 transmitter 202 and receiver 204 may be fabricatedas a monolithic integrated circuit. The only outboard componentsrequired for operation of the circuit are positive and negativereservoir capacitors 22 and 34 and the positive and negative transfercapacitors 10 and 24.

While the preferred embodiment of FIG. 7 shows a single RS-232transmitter 202 and a single RS-232 transmitter 204, those of ordinaryskill in the art will readily recognize that other combinations ofreceivers and transmitters could be added without departing from thespirit of the invention. It is noted, however, that an embodiment of thecircuit of FIG. 7 which contains only one or more RS-232 receivers 204,and no RS-232 transmitters 202, does not require a negative power supplyconnection. This is because the negative swing of the RS-232 formatsignal is usually disregarded by the receiver circuitry.

The RS-232 transmitter circuit 202, as well as the RS-232 receivercircuit 204 may be conventionally configured out of CMOS elements as iswell known in the art. For example, RS-232 transmitter circuit 202 maybe a CMOS inverter with a level shifter to translate TTL logic levels tothe RS-232 format, as is known in the art. Alternatively, it may beconfigured similarly to the MC 1488 circuit, manufactured by Motorola.RS-232 receiver circuits 204 may be a CMOS inverter with a level shifterto translate the incoming RS-232 format signal to TTL logic levels as isknown in the art. Alternatively, it may be configured similarly to theMC 1489 circuit manufactured by Motorola.

A preferred embodiment of the present invention has been disclosed.Those of ordinary skill in the art will readily recognize that otherembodiments are possible which do not differ in material respects. It isthe intention of the inventors to include such embodiment within thescope of the appended claims.

We claim:
 1. An integrated single-chip voltage supply comprising:a meansadapted to be connected to a supply voltage for charging a capacitorapproximately to a supply voltage; a means for coupling said chargedcapacitor in series with said supply voltage to create a source ofpositive voltage approximately double the supply voltage; and a meansfor inverting said doubled voltage and making it available on the samechip, thereby providing a source of two available supply voltages on thesame chip, one being a positive voltage approximately two times saidsupply voltage, and the other being a negative voltage approximately twotimes said supply voltage.
 2. The integrated single-chip voltage supplyof claim 1 further characterized by said coupling means including ameans for charging a second capacitor to approximately two times saidsupply voltage, and said inverting means including a means for invertingthe positive doubled voltage on said second capacitor to provide anavailable negative voltage on the same chip in magnitude approximatelytwice said supply voltage.
 3. An integrated voltage supply and RS-232transmitter circuit or the same semiconductor chip, comprising:an RS-232transmitter circuit; a means adapted to be connected to a supply voltagefor charging a capacitor approximately to a supply voltage; a means forcoupling said charged capacitor in series with said supply voltage tocreate a source of positive voltage approximately double the supplyvoltage; a means for inverting said doubled voltage and making itavailable on the same chip, thereby providing a source of two availablesupply voltages on the same chip, one being a positive voltageapproximately two times said supply voltage, and the other being anegative voltage approximately two times said supply voltage; and meansconnecting said doubled voltage and said inverted doubled voltage tosaid RS-232 transmitter circuit for powering said RS-232 transmittercircuit.
 4. The integrated single-chip voltage supply of claim 3 furthercharacterized by said coupling means including a means for charging asecond capacitor to approximately two times said supply voltage, andsaid inverting means including a means for inverting the positivedoubled voltage on said second capacitor to provide an availablenegative voltage of magnitude approximately twice said supply voltage.